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Wide VIN Dual Standard Buck Regulator With 3A/3A Continuous Output Current
ISL85033
The ISL85033 is a dual standard buck regulator capable of 3A per channel continuous output current. With an input range of 4.5V to 28V, it provides a high frequency power solution for a variety of point of load applications. The PWM controller in the ISL85033 drives an internal switching N-Channel power MOSFET and requires an external Schottky diode to generate the output voltage. The integrated power switch is optimized for excellent thermal performance up to 3A of output current. The PWM regulator switches at a default frequency of 500kHz and it can be user programmed or synchronized from 300kHz to 2MHz. The ISL85033 utilizes peak current mode control to provide flexibility in component selection and minimize solution size. The protection features include overcurrent, UVLO and thermal overload protection. The ISL85033 is available in a small 4mmx4mm Thin Quad Flat Pb-free (TQFN) package.
ISL85033
Features
* Wide Input Voltage Range from 4.5V to 28V * Adjustable Output Voltage with Continuous Output Current up to 3A * Current Mode Control * Adjustable Switching Frequency from 300kHz to 2MHz * Independent Power-Good Detection * Selectable In-Phase or Out-of-Phase PWM Operation * Independent, Sequential, Ratiometric or Absolute Tracking Between Outputs * Internal 2ms Soft-start Time * Overcurrent/Short Circuit Protection, Thermal Overload Protection, UVLO * Boot Undervoltage Detection * Pb-Free (RoHS Compliant)
Applications*(see page 24)
* General Purpose Point of Load DC/DC Power Conversion * Set-top Boxes * FPGA Power and STB Power * DVD and HDD Drives * LCD Panels, TV Power * Cable Modems
Related Literature*(see page 24)
* See AN1574 "ISL85033DUALEVAL1Z Wide VIN Dual Standard Buck Regulator With 3A/3A Output Current"
100 90 EFFICIENCY (%) 80 70 60 50 40 0.0 12VOUT 1MHz
0.5
1.0
1.5
2.0
2.5
3.0
OUTPUT LOAD (A)
FIGURE 1. EFFICIENCY vs LOAD, VIN = 28V, TA = +25C
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1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright Intersil Americas Inc. 2010. All Rights Reserved All other trademarks mentioned are the property of their respective owners.
ISL85033
Pin Configuration
ISL85033 (28 LD TQFN) TOP VIEW
SYNCOUT PGOOD1 PGOOD2 22 SYNCIN
28
27
26
25
SGND
NC
FS
24 23
COMP1 1 FB1 2 SS1 3 PGND1 4 BOOT1 5 PHASE1 6 PHASE1 7 8 VIN1 9 VIN1 10 EN1 11 VCC 12 13 VIN2 EN2 14 VIN2 PD
21 COMP2 20 FB2 19 SS2 18 PGND2 17 BOOT2 16 PHASE2 15 PHASE2
Pin Descriptions
PIN NUMBER 1, 21 2, 20 SYMBOL COMP1, COMP2 FB1, FB2 PIN DESCRIPTION COMP1/COMP2 is the output of the error amplifier. Feedback pin for the regulator. FB is the negative input to the voltage loop error amplifier. COMP is the output of the error amplifier. The output voltage is set by an external resistor divider connected to FB. In addition, the PWM regulator's power-good and undervoltage protection circuits use FB1/2 to monitor the regulator output voltage. Soft-Start pins for each controller. The SS1/2 pins control the soft-start and sequence of their respective outputs. A single capacitor from the SS pin to ground determines the output ramp rate. See the "Output Tracking and Sequencing" on page 16 for soft-start and output tracking/sequencing details. If SS pins are tied to VCC, an internal soft-start of 2ms will be used. Power ground connections. Connect directly to the system GND plane. Floating bootstrap supply pin for the power MOSFET gate driver. The bootstrap capacitor provides the necessary charge to turn on the internal N-Channel MOSFET. Connect an external capacitor from this pin to PHASE. Switch node output. It connects the source of the internal power MOSFET with the external output inductor and with the cathode of the external diode. The input supply for the power stage of the PWM regulator and the source for the internal linear regulator that provides bias for the IC. Place a minimum of 10F ceramic capacitance from each VIN to GND and close to the IC for decoupling. PWM controller's enable inputs. The PWM controllers are held off when the pin is pulled to ground. When the voltage on this pin rises above 2V, the PWM controller is enabled. Output of the internal 5V linear regulator. Decouple to PGND with a minimum of 4.7F ceramic capacitor. Synchronization output. Provides a signal that is the inverse of the SYNCIN signal.
3, 19
SS1, SS2
4, 18 5, 17
PGND1, PGND2 BOOT1, BOOT2
6, 7, 15, 16 8, 9, 13, 14
PHASE1, PHASE2 VIN1, VIN2
10, 12 11 23
EN1, EN2 VCC SYNCOUT
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Pin Descriptions (Continued)
PIN NUMBER 24 SYMBOL SYNCIN PIN DESCRIPTION Connect to an external signal for synchronization from 300kHz to 2MHz (negative edge trigger). SYNCIN is not allowed to be floating. When SYNCIN = logic 0, PHASE1 and PHASE2 are running at 180 out-of-phase. When SYNCIN = logic 1, PHASE1 and PHASE2 are running at 0 in-phase. When SYNCIN = an external clock, PHASE1 and PHASE2 are running at 180 out-of-phase. Set the internal switching frequency 20% lower than the external SYNC frequency applied to the SYNCIN pin. Signal ground connections. The exposed pad must be connected to SGND and soldered to the PCB. All voltage levels are measured with respect to this pin. This is a no connection pin. Frequency selection pin. Tie to VCC for 500kHz switching frequency. Connect a resistor to GND for adjustable frequency from 300kHz to 2MHz. Open drain power-good output that is pulled to ground when the output voltage is below regulation limits or during the soft-start interval. There is an internal 5M internal pull-up resistor. The exposed pad must be connected to the system GND plane with as many vias as possible for proper electrical and thermal performance.
25 26 27 22, 28
SGND NC FS PGOOD1, PGOOD2
-
PD
Typical Application Schematics
VOUT2 R5 25.5k C4 68pF COMP2 FB2 R6 8.06k C5 470pF R8 69.8k R2 8.06k C2 470pF FB1 R4 69.8k COMP1 R1 42.2k C1 68pF VOUT1
20 VCC VCC VCC FS 27 SS2 19 SS1 3 PGOOD2 VOUT2 3A L2 7H PGOOD1 PHASE2 15/16 C13 47F D2 B340B C12 10nF BOOT2 17 22 28
21
1
2 8/9 13/14 VIN1 VIN2 10F C72 L1 7H VOUT1 3A C71 20F
ISL85033
6/7 PHASE1 C8 10nF
5 BOOT1 4/18 24 23 SYNCIN SYNCOUT 12 26 EN2 NC 10 25 EN1 SGND VCC 11 4.7F
C9 D1 47F B340B
FIGURE 2. DUAL 3A OUTPUT (VIN RANGE FROM 4.5V TO 28V)
3
PGND1/2
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Typical Application Schematics (Continued)
VOUT1 FB2 R5 42.2k C4 68pF R7 0 FB2 C5 1nF COMP2 COMP1 R8 34k R6 8.06k COMP2 FB2 FB1 2 8/9 13/14 ISL85033 6/7 VIN1 VIN2 10F C72 PHASE1 C8 10nF L1 7H C9 VOUT1 6A C71 20F
20 VCC Css2 47nF FS SS2 SS1 Css1 47nF 27 19 3
21
1
PGOOD2 22 PGOOD1 28 PHASE2 15/16
VOUT1 C13 47F L2 7H D2 B340B
C12 10nF BOOT2 24 SYNCIN 23 SYNCOUT 4/18 17 12 EN2 26 NC 10 25 EN1 SGND VCC 11 4.7F
5
BOOT1
47F D1 B340B
FIGURE 3. SINGLE 6A OUTPUT (VIN RANGE FROM 4.5V TO 28V) CURRENT SHARING
4
PGND1/2
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Functional Block Diagram
PGOOD2 COMP2 BOOT2 FB2 + VOLTAGE MONITOR FAULT MONITOR CSA2 POWER-ON RESET MONITOR FAULT MONITOR 0.8V REFERENCE MONITOR VOLTAGE SS1 CONTROL SOFT-START -10% BOOT UV DETECTION VCC + EA COMP1 DRIVE GATE BOOT REFRESH CONTROL PGND1 + PHASE1 PGOOD1
VCC 5M -10% VCC BOOT UV DETECTION VIN2 CSA2 EA + COMP2 0.8V REFERENCE EN2 GATE DRIVE BOOT REFRESH CONTROL SLOPE COMP + PHASE2 PGND2 CSA2 THERMAL MONITOR +150C SYNCOUT OSCILLATOR FS SYNCIN SLOPE COMP + VIN1 CSA1 EN1 CSA1 BOOT1 SGND
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SS2
SOFT-START CONTROL + CSA1 FB1 +
VIN1
LDO
VCC = 5V VIN1
VCC
VCC
5M
5
COMP1 EPAD GND
ISL85033
Ordering Information
PART NUMBER (Notes 1, 2, 3) ISL85033IRTZ NOTES: 1. Add "-T*" suffix for Tape and Reel. Please refer to TB347 for details on reel specifications. 2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020. 3. For Moisture Sensitivity Level (MSL), please see device information page for ISL85033. For more information on MSL please see techbrief TB363. PART MARKING 850 33IRTZ TEMP. RANGE (C) -40 to +85 PACKAGE (Pb-Free) 28 Ld TQFN PKG. DWG. # L28.4x4
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Table of Contents
Pin Configuration ................................................................................................................................ 2 Pin Descriptions .................................................................................................................................. 2 Typical Application Schematics............................................................................................................ 3 Functional Block Diagram .................................................................................................................... 5 Ordering Information ......................................................................................................................... 6 Absolute Maximum Ratings ................................................................................................................ 8 Thermal Information .......................................................................................................................... 8 Recommended Operating Conditions .................................................................................................. 8 Electrical Specificaitons ..................................................................................................................... 8 Typical Performance Curves ............................................................................................................. .10 Detailed Description .......................................................................................................................... 16 Operation Initialization ..................................................................................................................... 16 Power-On Reset and Undervoltage Lockout ......................................................................................... Enable and Disable .......................................................................................................................... Power Good.................................................................................................................................... Output Voltage Selection.................................................................................................................. 16 16 16 16
Output Tracking and Sequencing ....................................................................................................... 16 Protection Features ........................................................................................................................... 17 Buck Regulator Overcurrent Protection ............................................................................................... 17 Thermal Overload Protection............................................................................................................. 18 BOOT Undervoltage Protection .......................................................................................................... 18 Application Guidelines ....................................................................................................................... 18 Operating Frequency ....................................................................................................................... Synchronization Control ................................................................................................................... Output Inductor Selection ................................................................................................................ Buck Regulator Output Capacitor Selection ......................................................................................... Current Sharing Configuration........................................................................................................... Input Capacitor Selection ................................................................................................................. Loop Compensation Design............................................................................................................... Theory of Compensation .................................................................................................................. PWM Comparator Gain Fm................................................................................................................ Power Stage Transfer Functions ........................................................................................................ Rectifier Selection ........................................................................................................................... Power Derating Characteristics.......................................................................................................... Layout Considerations...................................................................................................................... 18 18 18 18 19 19 19 20 20 20 21 22 22
Revision History ................................................................................................................................ 24 Products ............................................................................................................................................ 24 Package Outline Drawing ................................................................................................................. 25
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Absolute Maximum Ratings
VIN1/2 to GND. . . . . . . . . . . . . . . . . . . . . . -0.3V to +30V PHASE1/2 to GND . . . . . . . . . . . . . . . . . . . -0.3V to +33V BOOT1/2 to PHASE1/2 . . . . . . . . . . . . . . . . -0.3V to +5.5V FS to GND . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +5.5V SYNCIN to GND . . . . . . . . . . . . . . . . . . . . . -0.3V to +5.5V FB1/2 to GND . . . . . . . . . . . . . . . . . . . . . -0.3V to +2.95V EN1/2 to GND . . . . . . . . . . . . . . . . . . . . . . -0.3V to +5.5V PGOOD1/2 to GND . . . . . . . . . . . . . . . . . . . -0.3V to +5.5V COMP1/2 to GND . . . . . . . . . . . . . . . . . . . . -0.3V to +5.5V VCC to GND Short Maximum Duration . . . . . . . . . . . . . . 1s SYNCOUT to GND . . . . . . . . . . . . . . . . . . . . -0.3V to +5.5V SS1/2 to GND . . . . . . . . . . . . . . . . . . . . . . -0.3V to +5.5V ESD Rating Human Body Model (Tested per JESD22-A114) . . . . . . 3kV Charged Device Model (Tested per JESD22-C101E) . . 2.2kV Machine Model (Tested per JESD22-A115) . . . . . . . . 300V Latch Up (Tested per JESD-78A; Class 2, Level A) . . . 100mA
Thermal Information
Thermal Resistance JA (C/W) JC (C/W) QFN Package (Notes 4, 5) . . . . . . . 38 3 Maximum Junction Temperature (Plastic Package) . . +150C Maximum Storage Temperature Range . . . -65C to +150C Ambient Temperature Range . . . . . . . . . . . -40C to +85C Junction Temperature Range . . . . . . . . . . -55C to +150C Operating Temperature Range . . . . . . . . . . -40C to +85C Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . .see link below http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Recommended Operating Conditions
Temperature. . . . . . . . . . . . . . . . . . . . . . . -40C to +85C Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . 4.5V to 28V
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and result in failures not covered by warranty.
NOTES: 4. JA is measured in free air with the component mounted on a high effective thermal conductivity test board with "direct attach" features. See Tech Brief TB379 for details. 5. For JC, the "case temp" location is the center of the exposed metal pad on the package underside.
Electrical Specifications
TA = -40C to +85C, VIN = 4.5V to 28V, unless otherwise noted. Typical values are at TA = +25C. Boldface limits apply over the operating temperature range, -40C to +85C SYMBOL TEST CONDITIONS MIN (Note 8) TYP MAX (Note 8) UNITS
PARAMETER SUPPLY VOLTAGE VIN Voltage Range VIN Quiescent Supply Current VIN Shutdown Supply Current VCC Voltage POWER-ON RESET VIN POR Threshold
VIN IQ ISD VCC EN1/2 = 0V VIN = 12V; IOUT = 0mA Rising Edge Falling Edge
4.5 1.2 20 4.5 5.1
28 2.2 45 5.6
V mA A V
3.9 3.2 3.7
4.4
V V
OSCILLATOR Nominal Switching Frequency FSW FS = VCC Resistor from FS to GND = 383k Resistor from FS to GND = 40.2k FS Voltage Synchronization Frequency VFS FS = 100k SYNCIN = 600kHz 1.2MHz SYNCIN 4MHz Minimum Off-Time ERROR AMPLIFIER Error Amplifier Transconductance Gain FB1, FB2 Leakage Current Current Sense Amplifier Gain Reference Voltage RT gm VFB = 0.8V 0.18 0.792 125 205 10 0.21 0.8 285 100 0.24 0.808 A/V nA V/A V tOFF 600 130 780 420 500 300 2000 800 300 2000 820 580 kHz kHz kHz mV kHz kHz ns
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Electrical Specifications
TA = -40C to +85C, VIN = 4.5V to 28V, unless otherwise noted. Typical values are at TA = +25C. Boldface limits apply over the operating temperature range, -40C to +85C (Continued) SYMBOL TEST CONDITIONS SS1/2 = VDD ISS Rise Fall Percentage of the soft-start time ISINK = 3mA 82.5 MIN (Note 8) 1.5 1.4 TYP 2.5 2 MAX (Note 8) UNITS 3.5 2.6 ms A
PARAMETER Soft-Start Ramp Time Soft-Start Charging Current POWER-GOOD PG1, PG2 Trip Level PG to PGOOD1, PGOOD2 PG1, PG2 Propagation Delay PG1, PG2 Low Voltage ENABLE INPUT EN1, EN2 Leakage Current EN1, EN2 Input Threshold Voltage
91 85.5 10 100
94
% % %
300
mV
EN1/2 = 0V/5V Low Level Float Level High Level
-1
1 0.8
A V V V
1.0 2
1.4
SYNC INPUT/OUTPUT SYNCIN Input Threshold Falling Edge Rising Edge Hysteresis SYNCIN Leakage Current SYNCIN Pulse Width SYNCOUT Phase-shift to SYNCIN SYNCOUT Frequency Range SYNCOUT Output Voltage High SYNCOUT Output Voltage Low FAULT PROTECTION Thermal Shutdown Temperature TSD THYS Overcurrent Protection Threshold OCP Blanking Time POWER MOSFET Highside Internal BOOT1, BOOT2 Refresh Lowside PHASE Leakage Current PHASE Rise Time NOTES: 6. Test Condition: VIN = 28V, FB forced above regulation point (0.8V), no switching, and power MOSFET gate charging current not included. 7. Established by both current sense amplifier gain test and current sense amplifier output test @ IL = 0A. 8. Parameters with MIN and/or MAX limits are 100% tested at +25C, unless otherwise specified. Temperature limits established by characterization and are not production tested. tRISE RHDS RLDS IPHASE = 100mA IPHASE = 100mA EN1/2 = PHASE1/2 = 0V VIN = 25V 10 75 1 300 150 m nA ns Rising Threshold Hysteresis (Note 7) 4.1 150 20 5.1 60 6.1 C C A ns ISYNCOUT = 3mA Measured from rising edge to rising edge, if duty cycle is 50% 600 VCC - 0.3 VCC -0.08 0.08 0.3 SYNCIN = 0V/5V 100 180 4000 1.1 1.4 1.6 200 10 1000 1.9 V V mV nA ns kHz V V
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Typical Performance Curves
100 90 EFFICIENCY (%) 80 70 60 50 40 0.0 1.8VOUT 300kHz 0.5 1.0 1.5 2.0 OUTPUT LOAD (A) 2.5
Circuit of Figure 2. VIN = 12V, VOUT1 = 5V, VOUT2 = 3.3V, IOUT1 = 3A, IOUT2 = 3A, TA = -40C to +85C, unless otherwise noted. Typical values are at TA = +25C.
100 90 EFFICIENCY (%)
12VOUT 1MHz 9VOUT 1MHz 5VOUT 500kHz 3.3VOUT 500kHz
80 70 60 50 40 0.0 5VOUT 3.3VOUT
3.0
0.5
1.0
1.5
2.0
2.5
3.0
OUTPUT LOAD (A)
FIGURE 4. EFFICIENCY vs LOAD, TA = +25C, VIN = 28V
FIGURE 5. EFFICIENCY vs LOAD, TA = +25C, FSW = 500kHz, VIN = 12V
100 POWER DISSIPATION (W) 90 EFFICIENCY (%) 80 70 60 50 40 0 9VIN 12VIN 28VIN
4.2 3.5 2.8 2.1 1.4 0.7 0.0 28VIN 9VIN 0 1 2 3 4 OUTPUT LOAD (A) 5 6 12VIN
1
2 3 4 OUTPUT LOAD (A)
5
6
FIGURE 6. EFFICIENCY vs LOAD, TA = +25C, CURRENT SHARING 5VOUT, FSW = 500kHz
FIGURE 7. POWER DISSIPATION vs LOAD, TA = +25C, CURRENT SHARING 5VOUT, FSW = 500kHz
4.8 POWER DISSIPATION (W) OUTPUT VOLTAGE (V) 4.0 3.2 2.4 1.6 0.8 0.0 12VIN 28VIN 0 1 2 3 4 OUTPUT LOAD (A)
5.04 5.03 5.02 5.01 5.00 4.99 4.98 0 12VIN 9VIN 28VIN
9VIN 5 6
0.5
1.0 1.5 2.0 OUTPUT LOAD (A)
2.5
3.0
FIGURE 8. POWER DISSIPATION vs LOAD, TA = +85C, CURRENT SHARING 5VOUT, FSW = 500kHz
FIGURE 9. VOUT REGULATION vs LOAD, CHANNEL 1, TA = +25C, 5VOUT, FSW = 500kHz
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Typical Performance Curves
5.04 OUTPUT VOLTAGE (V)
Circuit of Figure 2. VIN = 12V, VOUT1 = 5V, VOUT2 = 3.3V, IOUT1 = 3A, IOUT2 = 3A, TA = -40C to +85C, unless otherwise noted. Typical values are at TA = +25C. (Continued)
3.329 OUTPUT VOLTAGE (V) 3.328 3.326 3.325 3.323 3.322 3.320 0 12VIN 0.5 1.0 1.5 2.0 2.5 3.0 28VIN 18VIN
5.03 5.02 5.01 5.00 4.99 4.98 0 28VIN 9VIN
12VIN
1
2
3
4
5
6
OUTPUT LOAD (A)
OUTPUT LOAD (A)
FIGURE 10. VOUT REGULATION vs LOAD, CURRENT SHARING, TA = +25C, 5VOUT, FSW = 500kHz
FIGURE 11. VOUT REGULATION vs LOAD, CHANNEL 2, TA = +25C, 3.3VOUT, FSW = 500kHz
5.04 OUTPUT VOLTAGE (V) OUTPUT VOLTAGE (V) 5.03 5.02 5.01 5.00 4.99 4.98 0 3A 2A 0A
5.02 5.01 5.00 4.99 4.98 4A 4.97 4.96 0 5 10 6A 0A
5
10
15
20
25
30
15
20
25
30
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
FIGURE 12. OUTPUT VOLTAGE REGULATION vs VIN, CHANNEL 1, TA = +25C, 5VOUT, FSW = 500kHz
FIGURE 13. OUTPUT VOLTAGE REGULATION vs VIN, CURRENT SHARING, TA = +25C, 5VOUT, FSW = 500kHz
3.340 OUTPUT VOLTAGE (V) 3.335 3.330 3.325 3.320 3.315 3.310 0 0A 2A 3A VOUT1 RIPPLE 20mV/DIV LX1 5V/DIV
IL1 0.1A/DIV 25 30
5
10
15
20
INPUT VOLTAGE (V)
FIGURE 14. OUTPUT VOLTAGE REGULATION vs VIN, CHANNEL 2, TA = +25C, 3.3VOUT, FSW = 500kHz
FIGURE 15. STEADY STATE OPERATION AT NO LOAD CHANNEL 1
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Typical Performance Curves
LX1 5V/DIV
Circuit of Figure 2. VIN = 12V, VOUT1 = 5V, VOUT2 = 3.3V, IOUT1 = 3A, IOUT2 = 3A, TA = -40C to +85C, unless otherwise noted. Typical values are at TA = +25C. (Continued)
LX2 5V/DIV VOUT1 RIPPLE 20mV/DIV VOUT2 RIPPLE 20mV/DIV IL1 0.2A/DIV IL2 0.1A/DIV
FIGURE 16. STEADY STATE OPERATION AT NO LOAD CHANNEL 1 (VIN = 9V)
FIGURE 17. STEADY STATE OPERATION AT NO LOAD CHANNEL 2
LX1 5V/DIV
LX2 5V/DIV
VOUT1 RIPPLE 20mV/DIV
IL1 1A/DIV
VOUT2 RIPPLE 20mV/DIV
IL2 1A/DIV
FIGURE 18. STEADY STATE OPERATION WITH FULL LOAD CHANNEL 1
FIGURE 19. STEADY STATE OPERATION WITH FULL LOAD CHANNEL 2
LX2 10V/DIV
VOUT RIPPLE 20mV/DIV
VOUT1 RIPPLE 20mV/DIV
LX1 10V/DIV
IL1 2A/DIV
FIGURE 20. STEADY STATE OPERATION WITH FULL LOAD CURRENT SHARING
FIGURE 21. LOAD TRANSIENT CHANNEL 1
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Typical Performance Curves
Circuit of Figure 2. VIN = 12V, VOUT1 = 5V, VOUT2 = 3.3V, IOUT1 = 3A, IOUT2 = 3A, TA = -40C to +85C, unless otherwise noted. Typical values are at TA = +25C. (Continued)
EN1 5V/DIV
VOUT1 2V/DIV VOUT2 RIPPLE 20mV/DIV
IL1 0.5A/DIV
PG1 5V/DIV
IL2 2A/DIV
FIGURE 22. LOAD TRANSIENT CHANNEL 2
FIGURE 23. SOFT-START WITH NO LOAD CHANNEL 1
EN2 5V/DIV VOUT2 2V/DIV
EN1 5V/DIV
VOUT1 2V/DIV IL1 2A/DIV
IL2 0.5A/DIV
PG2 5V/DIV
PG1 5V/DIV
FIGURE 24. SOFT-START WITH NO LOAD CHANNEL 2
FIGURE 25. SOFT-START AT FULL LOAD CHANNEL 1
EN2 5V/DIV VOUT2 2V/DIV EN1 5V/DIV
VOUT1 1V/DIV IL2 2A/DIV IL1 0.5A/DIV PG2 5V/DIV PG 5V/DIV
FIGURE 26. SOFT-START AT FULL LOAD CHANNEL 2
FIGURE 27. SOFT-DISCHARGE SHUTDOWN CHANNEL 1
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Typical Performance Curves
Circuit of Figure 2. VIN = 12V, VOUT1 = 5V, VOUT2 = 3.3V, IOUT1 = 3A, IOUT2 = 3A, TA = -40C to +85C, unless otherwise noted. Typical values are at TA = +25C. (Continued)
VOUT1 2V/DIV
EN2 5V/DIV
VOUT2 0.5V/DIV
VOUT2 2V/DIV EN1, 2 2V/DIV
IL2 0.5A/DIV
PG 5V/DIV
FIGURE 28. SOFT-DISCHARGE SHUTDOWN CHANNEL 2
FIGURE 29. INDEPENDENT START-UP SEQUENCING AT NO LOAD
VOUT1 2V/DIV
VOUT1 2V/DIV
VOUT2 2V/DIV
VOUT2 2V/DIV
EN1, 2 2V/DIV
EN1, 2 2V/DIV
FIGURE 30. RATIOMETRIC START-UP SEQUENCING AT NO LOAD
FIGURE 31. ABSOLUTE START-UP SEQUENCING AT NO LOAD
LX1 10V/DIV
LX1 10V/DIV
VOUT1 RIPPLE 20mV/DIV
VOUT2 RIPPLE 20mV/DIV
LX2 10V/DIV
LX2 10V/DIV
SYNC 5V/DIV
SYNC 5V/DIV
FIGURE 32. STEADY STATE OPERATION CHANNEL 1 AT FULL LOAD WITH SYNC FREQUENCY = 4MHz
FIGURE 33. STEADY STATE OPERATION CHANNEL 2 AT FULL LOAD WITH SYNC FREQUENCY = 4MHz
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Typical Performance Curves
Circuit of Figure 2. VIN = 12V, VOUT1 = 5V, VOUT2 = 3.3V, IOUT1 = 3A, IOUT2 = 3A, TA = -40C to +85C, unless otherwise noted. Typical values are at TA = +25C. (Continued)
PHASE1 10V/DIV
PHASE1 10V/DIV
IL1 2A/DIV VOUT1 2V/DIV
VOUT1 2V/DIV IL1 2A/DIV PG1 5V/DIV
PG1 5V/DIV
FIGURE 34. OUTPUT SHORT CIRCUIT CHANNEL 1
FIGURE 35. OUTPUT SHORT CIRCUIT HICCUP AND RECOVERY FOR CHANNEL 1
PHASE2 10V/DIV
PHASE2 10V/DIV
IL2 2A/DIV VOUT2 2V/DIV
VOUT2 2V/DIV
IL2 2A/DIV
PG2 5V/DIV
PG2 5V/DIV
FIGURE 36. OUTPUT SHORT CIRCUIT CHANNEL 2
FIGURE 37. OUTPUT SHORT CIRCUIT HICCUP AND RECOVERY FOR CHANNEL 2
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ISL85033
Detailed Description
The ISL85033 combines a standard buck PWM controller with an integrated switching MOSFET. The buck controller drives an internal N-Channel MOSFET and requires an external diode to deliver load current up to 3A. A Schottky diode is recommended for improved efficiency and performance over a standard diode. The standard buck regulator can operate from an unregulated DC source, such as a battery, with a voltage ranging from +4.5V to +28V. The converter output can be regulated to as low as 0.8V. These features make the ISL85033 ideally suited for FPGA, set-top boxes, LCD panels, DVD drives, and wireless chipset power applications. The ISL85033 employs a peak current mode control loop which simplifies feedback loop compensation and rejects input voltage variation. External feedback loop compensation allows flexibility in output filter component selection. The regulator switches at a default 500kHz and it can be adjusted from 300kHz to 2MHz with a resistor from FS to GND. The ISL85033 is also synchronizable from 300kHz to 2MHz.
voltage set by FB. When VOUT drops 10% below the nominal regulation voltage, the ISL85033 pulls PG low. Any fault condition forces PG low until the fault condition is cleared by attempts to soft-start. There is an internal 5M internal pull-up resistor.
Output Voltage Selection
The regulator output voltages is easily programmed using an external resistor divider to scale VOUT relative to the internal reference voltage. The scaled voltage is applied to the inverting input of the error amplifier; refer to Figure 38. The output voltage programming resistor, R2, depends on the value chosen for the feedback resistor, R3, and the desired output voltage, VOUT, of the regulator. Equation 1 describes the relationship between VOUT and resistor values. R3 is often chosen to be in the 1k to 10k range.
R 2 = ( V OUT - 0.8 ) * R 3 0.8 (EQ. 1)
Operation Initialization
The power-ON reset circuitry and enable inputs prevent false start-up of the PWM regulator output. Once all input criteria are met, the controller soft-starts the output voltage to the programmed level.
If the desired output voltage is 0.8V, then R3 is left unpopulated and R2 is zero ohm.
VOUT FB EA + R3 R2
Power-On Reset and Undervoltage Lockout
The ISL85033 automatically initializes upon receipt of input power supply. The power-on reset (POR) function continually monitors VIN1 voltage. While below the POR threshold, the controller inhibits switching of the internal power MOSFET. Once exceeded, the controller initializes the internal soft-start circuitry. If VIN1 supply drops below their falling POR threshold during soft-start or operation, the buck regulator is disabled until the input voltage returns.
0.8V REFERENCE
FIGURE 38. EXTERNAL RESISTOR DIVIDER
Output Tracking and Sequencing
Output tracking and sequencing between channels can be implemented by using the SS1 and SS2 pins. Figures 39, 40 and 41 show several configurations for output tracking/sequencing for a 2.5V and 1.8V application. Independent soft-start for each channel is shown in Figure 39 and measured in Figure 29. The output ramp-time for each channel (tSS) is set by the soft-start capacitor (CSS).
C SS [ F ] = 2.5*t SS ( s )
Enable and Disable
When EN1 and EN2 are pulled low, the device enters shutdown mode and the supply current drops to a typical value of 20A. All internal power devices are held in a high-impedance state while in shutdown mode. The EN pin enables the controller of the ISL85033. When the voltage on the EN pin exceeds its logic rising threshold, the controller initiates the 2ms soft-start function for the PWM regulator. If the voltage on the EN pin drops below the falling threshold, the buck regulator shuts down.
(EQ. 2)
Ratiometric tracking is achieved in Figure 40 by using the same value for the soft-start capacitor on each channel; it is measured in Figure 30. By connecting a feedback network from VOUT1 to the SS2 pin with the same ratio that sets VOUT2 voltage, absolute tracking shown in Figure 41 is implemented. The measurement is shown in Figure 31. If the output of Channel 1 is shorted to GND, it will enter overcurrent hiccup mode, SS2 will be pulled low through the added resistor between VOUT1 and SS2 and this will force Channel 2 into hiccup as well. If the output of Channel 2 is shorted to GND with VOUT1 in
Power Good
PG is the open-drain output of a window comparator that continuously monitors the buck regulator output voltage via the FB pin. PG is actively held low when EN is low and during the buck regulator soft-start period. After the soft-start period terminates, PG becomes high impedance as long as the output voltage (monitored on the FB pin) is above 90% of the nominal regulation
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regulation, it will enter overcurrent hiccup mode with a very short hiccup waiting time. The reason is that VOUT1 is still in regulation and can pull-up SS2 very quickly via the resistor added between VOUT1 and SS2. Figure 42 illustrates output sequencing. When EN1 is high and EN2 is floating, OUT1 comes up first and OUT2 won't start until OUT1 > 90% of its regulation point. If EN1 is floating and EN2 is high, OUT2 comes up first and OUT1 won't start until OUT2 > 90% of its regulation point. If EN1 = EN2 = high, OUT1 and OUT2 come up at the same time. Please refer to Table 1 for conditions related to Figure 42 (Output Sequencing).
TABLE 1. OUTPUT SEQUENCING EN1 High Floating High EN2 Floating High High VOUT1 First After VOUT2 > 90% Same time as VOUT2 VOUT2 After VOUT1 > 90% First Same time as VOUT1 Not Allowed
C1 0.1F SS2 C2 0.1F EN1 EN2 ISL85033 VOUT2 C4 3.3V VOUT1 C3 5.0V VOUT1 C3
SS1 C1 0.22F SS2
5.0V
ISL85033 VOUT2 C4
3.3V
R2 8.06k
R1 25.5k
NOTE FIGURE 41. ABSOLUTE START-UP
SS1
Floating Floating
SS1 C1 0.1F C2 0.22F
VOUT1 C3
5.0V
SS2 ISL85033 VOUT2 C4
3.3V
FIGURE 42. OUTPUT SEQUENCING
Protection Features
The ISL85033 limits the current in all on-chip power devices. Overcurrent protection limits the current on the two buck regulators and internal LDO for VCC.
FIGURE 39. INDEPENDENT START-UP
Buck Regulator Overcurrent Protection
VOUT1 C3 5.0V
SS1 C1 0.1F SS2
ISL85033 VOUT2 C2 0.1F C4
3.3V
During the PWM on-time, the current through the internal switching MOSFET is sampled and scaled through an internal pilot device. The sampled current is compared to a nominal 5A overcurrent limit. If the sampled current exceeds the overcurrent limit reference level, an internal overcurrent fault counter is set to 1 and an internal flag is set. The internal power MOSFET is immediately turned off and will not be turned on again until the next switching cycle. The protection circuitry continues to monitor the current and turns off the internal MOSFET as described. If the overcurrent condition persists for 17 sequential clock cycles, the overcurrent fault counter overflows indicating an overcurrent fault condition exists. The regulator is shut down and power-good goes low. The buck controller attempts to recover from the overcurrent condition after waiting 8 soft-start cycles.
FIGURE 40. RATIOMETRIC START-UP
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The internal overcurrent flag and counter are reset. A normal soft-start cycle is attempted and normal operation continues if the fault condition has cleared. If the overcurrent fault counter overflows during soft-start, the converter shuts down and this hiccup mode operation repeats. The falling edge on the SYNCIN triggers the rising edge of PHASE1/2. The switching frequency for each output is half of the SYNCIN frequency.
Output Inductor Selection
The inductor value determines the converter's ripple current. Choosing an inductor current requires a somewhat arbitrary choice of ripple current, I. A reasonable starting point is 30% of total load current. The inductor value can then be calculated using Equation 4:
L= VIN - VOUT Fs x I x VOUT VIN (EQ. 4)
Thermal Overload Protection
Thermal overload protection limits maximum junction temperature in the ISL85033. When the junction temperature (TJ) exceeds +150C, a thermal sensor sends a signal to the fault monitor. The fault monitor commands the buck regulator to shut down. When the junction temperature has decreased by 20C, the regulator will attempt a normal soft-start sequence and return to normal operation. For continuous operation, the +125C junction temperature rating should not be exceeded.
BOOT Undervoltage Protection
If the BOOT capacitor voltage falls below 2.5V, the BOOT undervoltage protection circuit will pull the phase pin low through a 1 switch for 400ns to recharge the capacitor. This operation may arise during long periods of no switching as in no load situations.
Increasing the value of inductance reduces the ripple current and thus ripple voltage. However, the larger inductance value may reduce the converter's response time to a load transient. The inductor current rating should be such that it will not saturate in overcurrent conditions.
Buck Regulator Output Capacitor Selection
An output capacitor is required to filter the inductor current. Output ripple voltage and transient response are 2 critical factors when considering output capacitance choice. The current mode control loop allows the usage of low ESR ceramic capacitors and thus smaller board layout. Electrolytic and polymer capacitors may also be used. Additional consideration applies to ceramic capacitors. While they offer excellent overall performance and reliability, the actual in-circuit capacitance must be considered. Ceramic capacitors are rated using large peakto-peak voltage swings and with no DC bias. In the DC/DC converter application, these conditions do not reflect reality. As a result, the actual capacitance may be considerably lower than the advertised value. Consult the manufacturers data sheet to determine the actual in-application capacitance. Most manufacturers publish capacitance vs DC bias so that this effect can be easily accommodated. The effects of AC voltage are not frequently published, but an assumption of ~20% further reduction will generally suffice. The result of these considerations can easily result in an effective capacitance 50% lower than the rated value. Nonetheless, they are a very good choice in many applications due to their reliability and extremely low ESR. The following equations allow calculation of the required capacitance to meet a desired ripple voltage level. Additional capacitance may be used. For the ceramic capacitors (low ESR): =
I V OUTripple = -------------------------------------8 F SW C OUT
Application Guidelines
Operating Frequency
The ISL85033 operates at a default switching frequency of 500kHz if FS is tied to VCC. Tie a resistor from FS to GND to program the switching frequency from 300kHz to 2MHz, as shown in Equation 3.
R FS [ k ] = 122k ( t - 0.17s ) (EQ. 3)
Where: t is the switching period in s.
300
RFS (k)
200
100
0 500 750 1000 1250 1500 1750 2000 FS (kHz)
(EQ. 5)
FIGURE 43. RFS SELECTION vs FS
Synchronization Control
The frequency of operation can be synchronized up to 2MHz by an external signal applied to the SYNCIN pin.
where I is the inductor's peak to peak ripple current, FSW is the switching frequency and COUT is the output capacitor. If using electrolytic capacitors then:
V OUTripple = I*ESR (EQ. 6)
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Regarding transient response needs, a good starting point is to determine the allowable overshoot in VOUT if the load is suddenly removed. In this case, energy stored in the inductor will be transferred to COUT causing its voltage to rise. After calculating capacitance required for both ripple and transient needs, choose the larger of the calculated values. The following equation determines the required output capacitor value in order to achieve a desired overshoot relative to the regulated voltage.
I OUT 2 * L C OUT = -------------------------------------------------------------------------------------------V OUT 2 * ( V OUTMAX V OUT ) 2 - 1 ) (EQ. 7)
IRMS/IO
If capacitors other than MLCC are used, attention must be paid to ripple and surge current ratings.
I RMS ----------- = Io D - D2 (EQ. 9)
where D = VO/VIN The input ripple current is graphically represented in Figure 45.
0.6 0.5 0.4 0.3 0.2 0.1 0
where VOUTMAX/VOUT is the relative maximum overshoot allowed during the removal of the load. For an overshoot of 5%, the equation becomes:
I OUT 2 * L C OUT = ---------------------------------------------------V OUT 2 * ( 1.05 2 - 1 ) (EQ. 8)
The graph in Figure 44 shows the relationship of COUT and % overshoot at 3 different output voltages. L is assumed to to be 7H and IOUT is 3A.
0
0.2
0.4 D
0.6
0.8
FIGURE 45. IRMS/IO vs DUTY CYCLE
80 COUT (F)
60 3.3VOUT 40 5VOUT 12VOUT
A minimum of 10F ceramic capacitance is required on each VIN pin. The capacitors must be as close to the IC as physically possible. Additional capacitance may be used.
Loop Compensation Design
ISL85033 uses a constant frequency current mode control architecture to achieve simplified loop compensation and fast loop transient response. The compensator schematic is shown in Figure 47. As mentioned in the COUT selection, ISL85033 allows the usage of low ESR output capacitor. Choice of the loop bandwidth fc is somewhat arbitrary but should not exceed 1/4 of the switching frequency. As a starting point, the lower of 100kHz or 1/6 of the switching frequency is reasonable. The following equations determine initial component values for the compensation, allowing the designer to make the selection with minimal effort. Further detail is provided in "Theory of Compensation" on page 20 to allow fine tuning of the compensator. Compensation resistor R1 is given by Equation 10:
2f c V o C o R T R 1 = ----------------------------------g m V FB (EQ. 10)
20
0 1.02
1.04
1.06 VOUTMAX/VOUT
1.08
1.10
FIGURE 44. COUT vs OVERSHOOT VOUTMAX/VOUT
Current Sharing Configuration
In current sharing configuration, FB1 is connected to FB2, EN1 to EN2, COMP1 to COMP2 and VOUT1 to VOUT2 as shown in Figure 3. As a result, the equivalent gm doubles its single channel value. Since the two channels are out-of-phase, the frequency will be 2X the channel switching frequency. Ripple current cancellation will reduce the ripple current seen by the output capacitors and thus lower the ripple voltage. This results in the ability to use less capacitance than would be required by a single phase design of similar rating. Ripple current cancellation also reduces the ripple current seen at the input capacitors.
which when applied to ISL85033 becomes:
R 1 [ k ] = 0.008247 f c V o C o (EQ. 11)
Input Capacitor Selection
To reduce the resulting input voltage ripple and to minimize EMI by forcing the very high frequency switching current into a tight local loop, an input capacitor is required. The input capacitor must have adequate ripple current rating which can be approximated by the Equation 9.
where Co is the output capacitor value [F], fc = loop bandwidth [kHz] and Vo is the output voltage [V]. Compensation capacitors C1 [nF], C2 [pF] are given by Equation 12:
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C o x V o x ( 10 ) C o x R c x ( 10 ) C 1 = ----------------------------------------- ,C 2 = ----------------------------------------Io x R1 R1
3 6
(EQ. 12)
CURRENT SAMPLING TRANSFER FUNCTION He(S) In current loop, the current signal is sampled every switching cycle. Equation 15 shows the transfer function:
S S H e ( S ) = ------ + -------------- + 1 2 Q nn n 2
2
where Io [A] is the output load current, R1 () and Rc () is the ESR of the output capacitor Co. Example: Vo = 5V, Io = 3A, fs = 500kHz, fc = 50kHz, Co = 47F/Rc = 5m, then the compensation resistance R1 = 96k. The compensation capacitors are: C1 = 815pF, C2 = 2.5pF (There is approximately 3pF parasitic capacitance from VCOMP to GND; therefore, C2 is optional).
(EQ. 15)
Where Qn and n are given by Q n = - -- , = n = f s .
Power Stage Transfer Functions
Transfer function F1(S) from control to output voltage is calculated in Equation 16:
S1 + ---------- esr vo F 1 ( S ) = ------ = V in ------------------------------------- 2 d S S ------ + -------------- + 1 2 Q op o 1 1 o Where esr = -------------- ,Q p R o ------ , o = -------------Rc Co L C LC o (EQ. 16)
Theory of Compensation
The sensed current signal is injected into the voltage loop to achieve current mode control to simplify the loop compensation design. The inductor is not considered as a state variable for current mode control and the system becomes a single order system. It is much easier to design a compensator to stabilize the voltage loop than voltage mode control. Figure 46 shows the small signal model of the synchronous buck regulator.
^ iIN ^ VIN + ^ ILd 1:D ^ iL L ^ VO + ^ VINd RT Rc Ro Co
Transfer function F2(S) from control to inductor current is given by Equation 17:
S 1 + ----- V in z Io F 2 ( S ) = ---- = -------------------- ------------------------------------- Ro + RL 2 d S S ------ + -------------- + 1 2 Q op o Where z = -------------- . Ro Co 1 (EQ. 17)
Current loop gain Ti(S) is expressed as Equation 18:
T i ( S ) = R T F m F 2 ( S )H e ( S )
(EQ. 18)
^ d Fm
Ti(S) K
The voltage loop gain with open current loop is calculated in Equation 19:
T v ( S ) = KFm F 1 ( S )A v ( S ) (EQ. 19)
+
He(S) ^ VCOMP -Av(S)
Tv(S)
The voltage loop gain with current loop closed is given by Equation 14:
Tv ( S ) L v ( S ) = ----------------------1 + Ti ( S ) V FB K = ---------- , V (EQ. 20)
FIGURE 46. SMALL SIGNAL MODEL OF SYNCHRONOUS BUCK REGULATOR
PWM Comparator Gain Fm
The PWM comparator gain Fm for peak current mode control is given by Equation 13:
1 d F m = ---------------- = ------------------------------ ( S e + S n )T s v comp (EQ. 13)
FB is the feedback voltage of the Where Vo voltage error amplifier. If Ti(S)>>1, then Equation 20 can be simplified as shown in Equation 21:
S 1 + ----------V FB R o + R L esr A v ( S ) 1 L v ( S ) = ---------- -------------------- --------------------- ---------------- , p -------------RT Vo Ro Co S- H ( S ) 1 + ------ e p
(EQ. 21)
Where Se is the slew rate of the slope compensation and Sn is given by Equation 14.
V in - V o S n = R t --------------------L (EQ. 14)
From Equation 21, it is shown that the system is a single order system, which has a single pole located at p before the half switching frequency. Therefore, a simple type II compensator can be easily used to stabilize the system.
Where: RT is trans-resistance, and is the product of the current sensing resistance and gain of the current amplifier in current loop.
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Vo
VFB = 0.8V, Se = 1.1x105V/s, Sn = 3.4x105V/s, fc = 80kHz, then compensator resistance R1 = 72k.
C3 V FB V REF GM + R1 C2 C1 V COMP
R2
Put the compensator zero at 6.6kHz (~1.5x CoRo), and put the compensator pole at ESR zero, which is 1.45MHz. The compensator capacitors are: C1 = 470pF, C2 = 3pF (There is approximately 3pF parasitic capacitance from VCOMP to GND; therefore, C2 is optional). Figure 48A shows the simulated voltage loop gain. It is shown that it has 80kHz loop bandwidth with 69 phase margin and 15dB gain margin. Optional addition phase boost can be added to the overall loop response by using C3.
60 45 30 GAIN (dB) 15 0 -15
R3
FIGURE 47. TYPE II COMPENSATOR
Figure 47 shows the type II compensator and its transfer function is expressed as Equation 22:
S S 1 + ------------ 1 + ------------ gm cz1 cz2 v comp ---------------- = -------------------- --------------------------------------------------------Av ( S ) = C1 + C2 S v FB S 1 + ---------
cp
(EQ. 22)
Where:
C1 + C2 1 1 cz1 = -------------- , cz2 = -------------- , cp = ---------------------R1 C1 C2 R1 C1 R2 C3 (EQ. 23)
-30
100
1*103
1*104
1*105
1*106
the compensator design goal is: High DC gain
Loop bandwidth fc: -- to ------ f s 4 10 1 1
100 80 60 40
FIGURE 48A.
Gain margin: >10dB Phase margin: 40 The compensator design procedure is shown in Equation 24:
1 Put compensator zero cz1 = ( 1to3 ) ---------------RO CO (EQ. 24)
PHASE ()
20 0 -20 100
Put one compensator pole at zero frequency to achieve high DC gain, and put another compensator pole at either ESR zero frequency or half switching frequency, whichever is lower. The loop gain Tv(S) at crossover frequency of fc has unity gain. Therefore, the compensator resistance R1 is determined by Equation 25:
2f c V o C o R T R 1 = ----------------------------------g m V FB (EQ. 25)
1*103
1*104
1*105
1*106
FIGURE 48B.
Rectifier Selection
Current circulates from ground to the junction of the external Schottky diode and the inductor when the highside switch is off. As a consequence, the polarity of the switching node is negative with respect to ground. This voltage is approximately -0.5V (a Schottky diode drop) during the off-time. The rectifier's rated reverse breakdown voltage must be at least equal to the maximum input voltage, preferably with a 20% derating factor. The power dissipation when the Schottky diode conducts is expressed in Equation 27:
V OUT P D [ W ] = I OUT V D 1 - --------------- V IN (EQ. 27)
where gm is the trans-conductance of the voltage error amplifier, typically 200uA/V. Compensator capacitor C1 is then given by Equation 26:
1 1 C 1 = ----------------- ,C 2 = -----------------------R 1 cz 2R 1 f esr (EQ. 26)
Example: VIN = 12V, Vo = 5V, Io = 3A, fs = 500kHz, Co = 220F/5m, L = 5.6H, gm = 200s, RT = 0.21,
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Where: VD is the voltage drop of the Schottky diode. Selection of the Schottky diode is critical in terms of the high temperature reverse bias leakage current which is very dependent on VIN and exponentially increasing with temperature. Due to the nature of reverse bias leakage vs temperature, the diode should be carefully selected to operate in the worst case circuit conditions. Catastrophic failure is possible if the diode chosen experiences thermal runaway at elevated temperatures. Please refer to Application Note for diode selection.
Layout Considerations
Layout is very important in high frequency switching converter design. With power devices switching efficiently between 100kHz and 600kHz, the resulting current transitions from one device to another cause voltage spikes across the interconnecting impedances and parasitic circuit elements. These voltage spikes can degrade efficiency, radiate noise into the circuit, and lead to device overvoltage stress. Careful component layout and printed circuit board design minimizes these voltage spikes. As an example, consider the turn-off transition of the upper MOSFET. Prior to turn-off, the MOSFET is carrying the full load current. During turn-off, current stops flowing in the MOSFET and is picked up by the Schottky diode. Any parasitic inductance in the switched current path generates a large voltage spike during the switching interval. Careful component selection, tight layout of the critical components, and short, wide traces minimizes the magnitude of voltage spikes. There are two sets of critical components in the ISL85033 switching converter. The switching components are the most critical because they switch large amounts of energy, and therefore tend to generate large amounts of noise. Next, are the small signal components which connect to sensitive nodes or supply critical bypass current and signal coupling. A multi-layer printed circuit board is recommended. Figure 50 shows the connections of the critical components in the converter. Note that capacitors CIN and COUT could each represent numerous physical capacitors. Dedicate one solid layer, usually a middle layer of the PC board, for a ground plane and make all critical component ground connections with vias to this layer. Dedicate another solid layer as a power plane and break this plane into smaller islands of common voltage levels. Keep the metal runs from the PHASE terminals to the output inductor short. The power plane should support the input power and output power nodes. Use copper filled polygons on the top and bottom circuit layers for the phase nodes. Use the remaining printed circuit layers for small signal wiring. In order to dissipate heat generated by the internal LDO and MOSFET, the ground pad should be connected to the internal ground plane through at least four vias. This allows the heat to move away from the IC and also ties the pad to the ground plane through a low impedance path. The switching components should be placed close to the ISL85033 first. Minimize the length of the connections between the input capacitors, CIN, and the power switches by placing them nearby. Position both the ceramic and bulk input capacitors as close to the upper MOSFET drain as possible. Position the output inductor and output capacitors between the upper and Schottky diode and the load.
Power Derating Characteristics
To prevent the ISL85033 from exceeding the maximum junction temperature, some thermal analysis is required. The temperature rise is given by Equation 28:
T RISE = ( PD ) ( JA ) (EQ. 28)
where PD is the power dissipated by the regulator and JA is the thermal resistance from the junction of the die to the ambient temperature. The junction temperature, TJ, is given by Equation 29:
T J = ( T A + T RISE )
(EQ. 29)
where TA is the ambient temperature. For the QFN package, the JA is +38C/W. The actual junction temperature should not exceed the absolute maximum junction temperature of +125C When considering the thermal design, remember to consider the thermal needs of the rectifier diode. The ISL85033 delivers full current at ambient temperatures up to +85C if the thermal impedance from the thermal pad maintains the junction temperature below the thermal shutdown level, depending on the Input Voltage/Output Voltage combination and the switching frequency. The device power dissipation must be reduced to maintain the junction temperature at or below the thermal shutdown level. Figure 49 illustrates the power derating versus ambient temperature for the ISL85033 EVAL kit. Note that the EVAL kit derating curve is based on total circuit dissipation, not IC dissipation alone.
120 110 100 90 80 70 60 50 40 30 20 10 0 0
MAXIMUM AMBIENT TEMPERATURE (C)
JA = 38C/W
1
2 3 4 5 6 7 8 9 10 11 12 ISL85033EVAL1ZB EVAL BOARD TOTAL POWER DISSIPATION (W)
FIGURE 49. POWER DERATING CURVE
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The critical small signal components include any bypass capacitors, feedback components, and compensation components. Place the PWM converter compensation components close to the FB and COMP pins. The feedback resistors should be located as close as possible to the FB pin with vias tied straight to the ground plane as required.
Fb1
Comp2
Comp1
Fb2
LX2 trace
Cboot
L1
LX1 trace
ISL85033 SL85033
D1
VOUT1 Cout1
Cboot
... ...
vias
L2
D2
Cout2 VOUT2 VOUT2
Cin1 Cin2
FIGURE 50. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS
23
VIN1 VIN2
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Revision History
The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to web to make sure you have the latest Rev. DATE 10/15/10 REVISION FN6676.2 CHANGE Added the following sentence to the "SYNCIN" description in the "Pin Descriptions" table on page 3: "Set the internal switching frequency 20% lower than the external SYNC frequency applied to the SYNCIN pin." Added the following sentence to "Synchronization Control" on page 18: "The switching frequency for each output is half of the SYNCIN frequency." Revised tape and reel note in "Ordering Information" on page 6 from: "Add "-T" suffix for Tape and Reel. Please refer to TB347 for details on reel specifications" to: "Add "-T*" suffix for Tape and Reel. Please refer to TB347 for details on reel specifications" This is in order to delineate all tape and reel options. 9/14/10 Corrected Eq. 1 on page 16 from: R 2 x0.8V R 3 = ---------------------------------V OUT - 0.8V to: R 2 = ( V OUT - 0.8 ) * R 3 0.8 Revised preceding paragraph from: "The output voltage programming resistor, R3, depends on the value chosen for the feedback resistor, R2, and the desired output voltage, VOUT, of the regulator. Equation 1 describes the relationship between VOUT and resistor values. R2 is often chosen to be in the 1k to 10k range." to: "The output voltage programming resistor, R2, depends on the value chosen for the feedback resistor, R3, and the desired output voltage, VOUT, of the regulator. Equation 1 describes the relationship between VOUT and resistor values. R3 is often chosen to be in the 1k to 10k range." 6/21/10 6/18/10 FN6676.1 FN6676.0 Changed MIN/MAX for "Soft-Start Charging Current" on page 9 from 1.5/2.5A to 1.4/2.6A Initial Release.
Products
Intersil Corporation is a leader in the design and manufacture of high-performance analog semiconductors. The Company's products address some of the industry's fastest growing markets, such as, flat panel displays, cell phones, handheld products, and notebooks. Intersil's product families address power management and analog signal processing functions. Go to www.intersil.com/products for a complete list of Intersil product families. *For a complete listing of Applications, Related Documentation and Related Parts, please see the respective device information page on intersil.com: ISL85033 To report errors or suggestions for this datasheet, please go to www.intersil.com/askourstaff FITs are available from our website at http://rel.intersil.com/reports/search.php
For additional products, see www.intersil.com/product_tree Intersil products are manufactured, assembled and tested utilizing ISO9000 quality systems as noted in the quality certifications found at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com 24
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Package Outline Drawing
L28.4x4
28 LEAD THIN QUAD FLAT NO-LEAD PLASTIC PACKAGE Rev 0, 9/06
4 . 00 PIN 1 INDEX AREA A 2 . 50 B 0 . 40 22 21 28 PIN #1 INDEX AREA CHAMFER 0 . 400 X 45
1 0 . 4 x 6 = 2.40 REF
4 . 00
2 . 50
0 . 40
15 0 . 10 2X 14 8
7
0 . 20 0 . 05
TOP VIEW
0 . 4 x 6 = 2 . 40 REF 3 . 20
0 . 10 M C A B
BOTTOM VIEW
SEE DETAIL X'' (3 . 20) PACKAGE BOUNDARY (28X 0 . 20) MAX. 0 . 80 SEATING PLANE 0 . 00 - 0 . 05 0 . 20 REF 0 . 08C 0 . 10C C
SIDE VIEW
(2 . 50) (3 . 20)
(0 . 40)
C
0 . 20 REF
5
(0 . 40) (2 . 50) (28X 0 . 60)
0 ~ 0 . 05
DETAIL "X" TYPICAL RECOMMENDED LAND PATTERN
NOTES: 1. Controlling dimensions are in mm. Dimensions in ( ) for reference only. 2. Unless otherwise specified, tolerance : Decimal 0.05 Angular 2 3. Dimensioning and tolerancing conform to AMSE Y14.5M-1994. 4. Bottom side Pin#1 ID is diepad chamfer as shown. 5. Tiebar shown (if present) is a non-functional feature.
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3 . 20

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